The term “power supply” typically refers to a circuit that converts the electrical utility's AC mains to a DC voltage of selected and regulated value for use by some circuit or mechanism. A bench power supply is generally considered to be an item of electronic test equipment that replaces (temporarily) a built-in power supply for some product or apparatus under development. While a bench power supply might be considered a thing of a more general nature than a built-in power supply, in that it might be expected to operate under quite varied and possibly extreme conditions, the design of any power supply that is capable of supplying high currents raises certain issues common to power supplies in general. For example, the limiting of inrush current for certain vulnerable components within the power supply, such as the rectifier (often a full wave bridge) and the bulk filter capacitance ahead of a series pass regulator, is often a concern, as those components can undergo catastrophic failure under over-current conditions. Such over-current conditions can occur for initial turn on or for rectified cycle by cycle maintenance of a heavy load, and are always items of interest for the designer of the power supply. Another area of concern is noise. Quite aside from any necessary fans for cooling, the hum caused by the power transformer that (for a low voltage power supply) steps the AC mains down to a lower voltage often produces an audible hum that becomes more pronounced as the load on the supply increases.
While various things might be done to reduce the amount of hum, there is one trend in modern power supply design that, in the name of efficiency (always a good thing) can make a mere hum into an aggravating noise: switching elements in a power supply can cause large excursions in current that don't necessarily begin or end at a zero crossing and that can have very abrupt starts and stops. Since the power supplied by the power supply is at one point a magnetic force within the power transformer, its mechanical nature must ‘anchor’ (conduct while containing) the magnetic forces involved (or at least resist mechanical deformation or movement under their influence) to efficiently transform one AC voltage and current to another. Loose laminations within the magnetic core of the power transformer, and limitations on how tightly windings can be affixed thereto, provide the causes for the familiar hum at the power line frequency. These same mechanical conditions can turn that hum into more of a howl under the stimulation of the large and abrupt variations in current required by some switching techniques. This acoustic ‘switching noise’ noise from a switching power supply can sometimes be very objectionable. Abrupt variations in switched non-sinusoidal currents can also cause undesirable electrical noise in nearby components or circuits.
One technique that has been used to both provide filtering and a limit on inrush current is an initial series inductance in the path of the current. When provided as a separate individual component this is known as a ‘choke input’ filter. It can store power as a magnetic field in its winding and contribute a sustaining EMF (Electro Motive Force) and an associated current as a rectified cycle loses amplitude. It is not that this does not work, but the additional separate inductor is heavy, takes space, is expensive, adds its own acoustic noise, and is generally more effective as the load increases. All in all, it is no longer a first choice in many applications. There is, however, a close relative that can be used to advantage to limit inrush current: ‘leakage inductance’ within the power transformer itself.
Leakage inductance is not something that has oozed out a an inductor that is leaky, and that is lost in a puddle somewhere. It is real inductance that a transformer winding exhibits, according to the extent that the transformer fails in a certain way to behave as an ideal transformer. Consider identical primary and secondary windings on a perfect magnetic core having no losses and equally responsive to all of the magnetic flux created by each winding, and let each winding be completely coupled to all of the flux in the core. Let's operate this arrangement at low frequencies that pose no RF (Radio Frequency) related issues. Under these circumstances we have an ideal 1:1 transformer. We put 120 VAC at 60 Hz in and we expect (and do) get 120 VAC out. In fact, (and ignoring such artifacts as phase inversion) so long as we don't ask it to supply too much power to a load, we are hard pressed to tell by looking at the output voltage that it is coming from a transformer. There will, of course, come a point (for a non-ideal transformer) with increasing load that causes saturation in the core, and that behavior will probably betray the source as being a transformer.
That thought leads to another: How is it that this 1:1 transformer ‘knows’ to cause in its secondary winding a voltage that will produce the same current that is entering and leaving the primary winding? Worse still, suppose it was 5:1? And perhaps worst of all, how is it that the current in the primary falls to (very nearly) zero when the load is disconnected from the secondary? Just what is it that makes a transformer with no moving parts, and no on-board microprocessor, so smart?
The answer, and it has been known for a long time, is that transformer action is dependent upon a balance and counter balance of intervening effects that exactly cancel each other. The varying magnetic field of the primary induces a current in the secondary that in turn produces its own magnetic field that is then experienced by the primary. The two magnetic fields do not annihilate each other (or we'd have no transformer . . . ) but they do remove from each winding what it would, it were the only winding on the core, see as (self) inductance. So, each winding and its current has, courtesy of the other winding and the other current, ‘no’ inductive reactance to limit its current flow at the frequency of the applied EMF. The current in the secondary is limited by some (let us say resistive) external load. That sets the amount of flux that the secondary has available to influence the primary, and that limited influence allows the current in the primary to rise to the level needed to create the balance. The balance is, within limits, stable, and we call this machine with ‘no moving parts’ a transformer. If the load on the secondary is removed, there is no current in the secondary, and it creates no flux to influence the primary winding, whose current is then limited by just the ‘magnetizing inductance’ available in the primary, which might be considerable. The primary current is then very low, owing to an apparent inductive reactance of the primary.
This happy state of affairs relies on the idea that all of the lines of magnetic flux created by the primary are ‘consumed by’ (coupled to) the primary, and vice versa. Now suppose that, either because we treat certain secondary windings a certain way, or because of less than ideal magnetic coupling of each winding to the core (and mindful that magnetic coupling is ‘two way,’ or from both ‘from’ and ‘to’), some magnetic flux is not coupled from one winding to the other. Suppose it ‘leaks’ out of the core and then ‘back in’ without having intersected the other (primary) winding. (Or suppose some flux from a winding never enters the core and thus does not enter the other winding.) That magnetic flux from the one winding that ‘leaked out’ to not couple to the other is free to make inductive reactance in that first winding, since it does not participate in the balancing action for transformer operation described above. We have leakage flux giving rise to self induction. It is what we call leakage inductance. If we were to go back to our claim that one could not tell that there was a one to one transformer in a certain set-up, we now must admit that if it had any significant leakage inductance we could spot it as readily as we could the presence of a series inductor separate from the transformer. That is, we could detect a phase shift between the voltage at the terminals of the secondary winding and the current that it supplies.
This is not necessarily a bad thing, and the presence of leakage inductance in power transformers has successfully been used to replace a separate filter inductor in the input to the balance of the power supply. The right amount of such inductance can do good things besides mere filtering, such as improve the power factor that the whole power supply presents to the AC mains, and significantly reduce inrush currents.
However, where there is inductive reactance, there is apt to be some phase shift, and in this case it will increase with increasing current. (Think of a fixed L driven by an AC source and in series with a decreasing R to ground. As the R decreases the phase shift between the applied EMF and the drop across that resistor increases until it reaches 90° when R=0 and the current is limited solely by the reactance of the L.)
Switching power supplies do what they do using an appreciation of zero crossings in applied AC voltages. Significant phase shift ahead of a zero crossing detector in a switching power supply is by itself an unwelcome development, and has in the past been the subject of amelioration through a fixed amount of compensation. That has sometimes been an agreeable solution, but in high current conventional switching power supply applications the whole notion of leakage inductance as a useful current limiting device comes unstuck. The amount of leaking inductance is not constant, but is a non linear function of current, and the thus unpredictable phase shift that accompanies it becomes a source of instability in the control loop that operates the switching mechanism for voltage regulation within the internal operation of the supply (say, for a switching pre-regulator ahead of a final linear series pass regulator). We should like to actually increase the amount of leakage inductance in the power transformer for a high current power supply (in this case, a high performance bench power supply—but it might be for any power supply) to assist in limiting inrush currents and provide filtering, but are faced with an uncooperative conventional control loop architecture, such as the one shown in FIG. 1.
Briefly, FIG. 1A shows a switching power supply 1 wherein an AC source (‘the line voltage’) 2 is coupled through a suitable switching arrangement 3 and fusing arrangement to the primary winding 5 of a power transformer 4. A main secondary winding 6 supplies the main power provided by the power supply 1 to a load 22. In pursuit of this, a full wave bridge rectifier 10 creates from the main secondary's voltage 15 a full wave rectified voltage that is applied to a bulk filter capacitor 19 (which is a rather large value of capacitance) through a switching mechanism 18. The purpose of the switching mechanism 18 is to provide only the amount of charge to the bulk filter capacitor needed to allow a linear voltage/current regulator 20 to operate properly, while also limiting the amount of power that is dissipated by that voltage/current regulator (which may be a series pass arrangement). The idea is that, for light loads and/or lower output voltages the bulk capacitor might need a lesser charge, and the lower voltage associated with that reduces power dissipation. Regulation is obtained by monitoring (21) the voltage drop Va-Vb (23) across the linear regulator 20 (which we assume knows what the output voltage is to be, and have not shown its internal process to maintain it . . . ) and applying that voltage drop 23 to a controller 16 that also receives zero crossing information 13. The controller 16 creates a switch control signal 17 that causes the switching operation of the switching element 18, which might be a transistor (either a FET [Field Effect Transistor] or perhaps bi-polar transistor). It will be noted that the main secondary winding 6 is shown as exhibiting a desired intrinsic, or leakage, inductance 8. This is there on purpose to assist in limiting inrush current and provide some (slight) filtering.
An auxiliary secondary winding 7 is used to provide other auxiliary voltages, which may include ‘housekeeping’ voltages that are used in the operation of the main power supply circuitry for the main secondary winding (6). To this end, a full wave bridge 11 provides from the auxiliary secondary's voltage 14 a full wave rectified voltage that is used in two ways. First, it supplies a rectified voltage to the auxiliary voltage regulator circuitry 24; and second, it feeds into a zero crossing detector 12 whose output 13 is applied to controller 16. We have two things to note about this arrangement.
First, secondary winding 7 also exhibits a leakage inductance 9 that, to a certain degree, will mimic that of leakage inductance 8 in the main secondary winding. This is an attempt to provide zero crossing information that matches what is actually happening for the secondary 6. Unfortunately, it will only be an approximation, because of the untidy nature of leakage inductance 8, and the fact that the current in the two windings 6 and 7 do not react to their respective leakage inductances 8 and 9 in identical ways. Second, the full wave rectified waveform applied to the zero crossing detector 12 is not an agreeable one where there is a zero crossing every 180° of the applied source from the mains. We are not saying its frequency has changed, but rather that alternate ones of the zero crossings (every other zero crossing of the full wave rectified signal) has shifted to later in time by as much as 30° of the entire 360° of the complete AC input sine wave. This shift in every other zero crossing location is cause by the leakage inductance 8 in conjunction with non sinusoidal current wavefroms, and owing to mutual coupling within the power transformer also manifests itself to a significant degree in the auxiliary secondary winding 7. This ‘staggered’ zero crossing situation complicates the task that the zero crossing detector 12 must perform (we shall see that it job is to make a constant amplitude ramp in phase with the zero crossings).
A further disadvantage of the technique of FIG. 1A is shown in FIG. 1B, to which we now turn. Waveform diagram 25 in FIG. 1B shows some voltage and current relationships of interest in appreciating the operation of the circuit of FIG. 1A. In particular, note the phase shift 26 between auxiliary secondary (7) S2 and the main secondary (6) S1, and the generally asymmetrical nature of the zero crossings and their respective half cycles 30 and 31. Note the S1 current waveform (which primarily charges capacitor 19) has an abrupt trailing edge 28 that aligns with the S2 zero crossing 29. This is the occasion for acoustic noise from the power transformer and other aggravations (e.g., unwelcome electrical noise in the surrounding circuits that is of a significant amount).
We should like to keep the general outline of the switching pre-regulator (18) and final regulator (20) technique of FIG. 1A, but fix its assorted problems in a way that accommodates a larger than usual amount of leakage inductance 8 for use with relatively high currents, reduces acoustic and electrical noise from the power transformer when under load, and avoids instability in the pre-regulator's control loop arising from the control signal 13 exhibiting artifacts arising from the asymmetrical half-cycles (30, 31) from secondary winding S2 and its rectifier. What to do?